Current driver for diode laser system

ABSTRACT

An architecture for current driver circuitry for diode laser systems is contemplated whereby the circuitry is both modular and minimally complex with respect to the number of components and connections.

CROSS-REFERENCE TO RELATED APPLICATION

This application claims priority to U.S. Prov. Pat. App. Ser. No.62/083,787, filed 24 Nov. 2014 and entitled EFFICIENT HIGH SPEED HIGHPOWER CURRENT DRIVER FOR DIODES, the entirety of which is herebyincorporated by reference for all intents and purposes.

BACKGROUND

Suppliers of laser solution products for commercial and scientificapplications typically encounter a number of obstacles in the effort tobring a particular product to market. For example, the investmentrequired to develop new implementation-specific circuitry may in manyinstances be prohibitively expensive. To address such and other issues,embodiments of an architecture for current driver circuitry for diodelaser systems are contemplated whereby the circuitry is both modular andminimally complex with respect to the number of components andconnections. By extension, the circuitry may be less prone to failure,and a significant savings may be realized both in terms of systemdevelopment and maintenance.

SUMMARY

Embodiments of an architecture for current driver circuitry for diodelaser systems are shown and described whereby the circuitry is bothmodular and minimally complex with respect to the number of componentsand connections. The circuitry is modular because the same circuitry maybe used to drive many different diode loads with as little as changingthe magnitude of voltage bias to shift the operating point of thecircuitry. The architecture of the circuitry does not need to change.For example, some or all components of the circuitry may exhibit acompliance voltage of 60V or less, which is advantageous becausehigh-voltage PCB board certification or verification is not required,and the circuitry may regulate at or more than 200 A through a diodeload and be fast-modulating (e.g., 0-100 kHz, 0-100% duty cycle). A DCoffset voltage may be leveraged to compensate for the turn-on or kneevoltage of the diode load, and as such the circuitry may driveadditional diodes in series with the diode load at 100V, for example, atthe same current level even with a change in diode load and according tospecification. This is despite the 100V voltage level being greater thanthe 60V compliance voltage of the circuitry. Additionally, a powerscaling aspect may be realized by connecting in parallel identicalmodules of the current driver circuitry. Further, the circuitry may beminimally complex with respect to the number of components andconnections because a particular component of the circuitry may berealized with a single on-board PCB circuit component or element. Thepower supply that provides the DC offset voltage may not be an on-boardcircuit component or element. The same is true of the diode load. It iscontemplated that the circuitry may be realized in other ways, wherebyother circuitry may be built into the current driver circuitry. Whileadditional development and component costs would be incurred andadditional failure mechanisms may manifest, in some circumstances, suchalternatives may be beneficial.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a diode laser system according to thedisclosure.

FIG. 2 is a first circuit diagram of the system shown in FIG. 1.

FIG. 3 is a second circuit diagram of the system shown in FIG. 1.

FIG. 4 is a chart diagram of a first I-V characteristic according to thedisclosure.

FIG. 5 is a chart diagram of a second I-V characteristic according tothe disclosure.

FIG. 6 is a circuit diagram of a current regulator according to thedisclosure.

FIG. 7 is a chart diagram of switching responses of the regulator shownin FIG. 6.

FIG. 8 is a block diagram of a power scaling aspect according to thedisclosure.

DETAILED DESCRIPTION

Embodiments of an architecture for current driver circuitry for diodelaser systems is shown and described whereby the circuitry is bothmodular and minimally complex with respect to the number of componentsand connections. By extension, the circuitry may be less prone tofailure, and a significant savings may be realized both in terms ofsystem development and maintenance. Although not so limited, anappreciation of the various aspects of the present disclosure may begained from the following discussion in connection with the drawings.

For example, FIG. 1 shows a block diagram of an embodiment of a diodelaser system 100, equivalently, laser diode system 100, according to thedisclosure. In general, the diode laser system 100 is a kilowatt-classlaser system for materials processing and industrial applications (e.g.,for optical pumping of solid state lasers). One of skill in the art willhowever appreciate that the diode laser system 100 as described belowmay be leveraged for other applications, such as low-power LED-driverapplications for example. Additionally, one of skill in the art willappreciate that the diode laser system 100 as described below mayinclude a number of elements or components other than that shown. Forbrevity, however, a discussion directed to such elements or componentsis omitted.

The diode laser system 100 includes a power supply 102, a current driver104, and a laser diode load 106. In some embodiments, the current driver104 alone is built on a low-voltage rated circuit board and thus nocomponent of the current driver 104 has a compliance voltage of greaterthan 60V. This is advantageous because such a circuit board would not besubject to high-voltage certification or verification. The power supply102 includes an AC/DC converter 108, the current driver 104 includes acurrent regulator 110 and a switch element 112, and the diode load 106includes a diode stack 114. The diode stack 114 includes a string oflaser diodes, such as two series-connected stacks of twelveseries-connected laser diodes each for example. In this example, thediode stack 114 has a turn-on voltage of about 45.6V assuming that eachone of the twenty-four series-connected laser diodes has a turn-onvoltage of about 1.9V. These values are used for example purposes only.Other examples are possible.

In operation, the AC/DC converter 108 generates from an AC input voltage116 (alternatively, a DC input voltage may be leveraged) a first DCsupply voltage (VDC1) 118 and a second DC supply voltage (VDC2) 120 topower the current driver 104. The current driver 104 in turn generates adrive current (IDC) 122 to drive the diode load 106. A value ormagnitude of the drive current 122 is programmable and is defined by, oris a function of, a drive current setpoint 124. A laser output 126 isformed by combining optical emissions from each one of the plurality ofseries-connected laser diodes of the diode stack 114.

The first DC supply voltage 118 also biases the switch element 112 andthe operating point of the current regulator 110 to enable the currentdriver 104 to drive the diode stack 114, even though the turn-on or kneevoltage of the diode stack 114 (e.g., VK=45.6V) is greater than thecompliance voltage of the current regulator 110 (e.g., VC=41.5V). Thecompliance voltage of the current regulator 110 is the maximum voltagethe current regulator 110 will output in attempt to source a programmedcurrent (e.g., drive current 122) under normal operating conditions.Similarly, the compliance voltage range of the current regulator 110 isthe range of voltage values the current regulator 110 will output inattempt to source a programmed current (e.g., drive current 122) undernormal operating conditions. In other words, within its (compliancevoltage range) limits the current regulator 110 can maintain a constant(programmed) current output.

The magnitude of the compliance voltage range of the current regulator110 is defined by the compliance voltage of the current regulator 110.For example, when the compliance voltage of the current regulator 110 is60V, the drive voltage level may range from 0V-60V (inclusive), or from60V-120V (inclusive), or from 120V-180V (inclusive), while the currentregulator 110 remains within its compliance voltage range of 0V-60V. Afirst and second example modular and minimally complex realization ofthe diode laser system 100, and particularly the current driver 104 ofthe diode laser system 100, is shown and described in connection withFIG. 2 and FIG. 3.

FIG. 2 shows a first circuit diagram 200 of the diode laser system 100of FIG. 1. FIG. 3 shows a second circuit diagram 300 of the diode lasersystem 100 of FIG. 1. As shown in FIG. 2, the switch element 112 of thecurrent driver 104 is realized as a high-power, high-gain PMOStransistor. As shown in FIG. 3, the switch element 112 of the currentdriver 104 is realized as a high-power, high-gain NMOS transistor. It iscontemplated that an NMOS transistor may be incorporated into thearchitecture of the first circuit diagram 200, instead of the PMOStransistor. In such an embodiment, however, additional circuitry wouldbe built around the NMOS transistor (e.g., to measure drive current,enable/disable the transistor, etc.), and thus the architecture of thefirst circuit diagram 200 would be more complex than that as shown inboth FIG. 2 and FIG. 3.

Additionally, one of skill in the art will appreciate that thearchitecture of the first circuit diagram 200 is an inverted version ofthe architecture of the second circuit diagram 300, and vice versa.Thus, even though the features or aspects of the present disclosure arediscussed herein with reference to the first circuit diagram 200 of FIG.2, the same or similar principle of operation of the circuit illustratedby the first circuit diagram 200 is applicable to the circuitillustrated by second circuit diagram 300 of FIG. 3. In practice,however, the current driver 104 of the first circuit diagram 200 is ingeneral a current source, whereas the current driver 104 of the secondcircuit diagram 300 is in general a current sink.

With reference to FIG. 2, the AC/DC converter 108 generates VDC1 andVDC2 from the AC input voltage 116. Each one of VDC1 and VDC2 has afixed, regulated DC voltage, such as VDC1=VDC2=41.5V for example. Otherexamples are however possible. For instance, it may be advantageous toleverage one or more commodity power supplies that exhibit a supplyvoltage of 24V, 48V, 54V, and etc., since such power supplies arerelatively inexpensive and in general are widely used and available.Additionally, while two batteries or supply voltages are shown in FIG.2, other configurations or arrangements are possible. For example, threeor more batteries or supply voltages may be leveraged, such as threebatteries or supply voltages each of 48V, whereby VDC1=96V andVDC2=VDC1+48V=144V. Such a configuration may be advantageous when, forexample, the diode stack 114 includes three series-connected diodestacks in which each diode stack has a turn-on voltage of 40V so thatthe turn-on voltage of the diode stack 114 is 120V. In this and othersimilar examples, the current driver 104, and more specifically thecurrent regulator 110, may drive the diode stack 114 even though thecompliance voltage of the current regulator 110 is less than the turn-onvoltage of the diode stack 114.

For example, and as shown in FIG. 2, VDC1 and VDC2 are connected inseries whereby VDC1 and (VDC1+VDC2) are provided as the bottom powerrail and the top power rail of the current regulator 110, respectively.Voltage levels VDC1 and (VDC1+VDC2) are with respect to the common nodeshared by the cathodes of the DC supply 118 and the diode stack 114.Advantageously, such a configuration effectively biases or shifts theoperating point of the current regulator 110 to enable the currentregulator 110 to fully drive the diode stack 114, even though theturn-on voltage of the diode stack 114 (e.g., VK=45.6V) is greater thanthe compliance voltage of the current regulator 110 (e.g., VC=41.5V).This is further illustrated in FIG. 4.

FIG. 4 shows a chart diagram 400 of certain voltage mappings of thediode laser system 100 of the present disclosure. In particular, a diodecharacteristic 402 as shown in FIG. 4 corresponds to the I-Vcharacteristic of the diode stack 114, whereby the voltage VKcorresponds to the turn-on voltage of the diode stack 114 (e.g.,VK=45.6V). The turn-on voltage of the diode stack 114, along with thevoltage range that corresponds to the entire high-current part of thediode characteristic 402, falls within a voltage range that correspondsto a normal operating range of the current regulator 110. This isbecause, as mentioned above, (VDC1+VDC2) corresponds to the top powerrail of the current regulator 110 (e.g., VDC1+VDC2=41.5V+41.5V=83V), andVDC1 corresponds to the bottom power rail of the current regulator 110(e.g., VDC1=41.5V). One of skill in the art will appreciate that, sincethe current regulator 110 is operating closed-loop, the maximum andminimum drive voltages output by the current regulator 110 during normaloperation correspond to the top power rail and the bottom power rail ofthe current regulator 110, respectively. In this manner, the supplyvoltage configuration as shown in FIG. 2 (and FIG. 3) effectively biasesor shifts the operating point of the current regulator 110 to enable thecurrent regulator 110 to fully drive the diode stack 114, even thoughthe turn-on voltage of the diode stack 114 is greater than thecompliance voltage of the current regulator 110.

When connected as shown in FIG. 2 (or FIG. 3), the PMOS transistor 112(or NMOS transistor 112 of FIG. 3) is an active component that monitorsor detects the drive voltage that is output by the current regulator 110and responds accordingly. More specifically, the PMOS transistor 112 asconnected as shown in FIG. 2 is self-regulating and thus may beconsidered a sensor that changes its state in response to particularstimulus. For example, when the drive voltage level of the currentregulator 110 exceeds the knee or turn-on voltage of the diode stack114, VSG is positive. The PMOS transistor 112 is “on” because the gatenode of the PMOS transistor 112 is tied to VDC1 (e.g.,VSG>VK−VDC1=45.6V-41.5V=4.1V) and VSG exceeds the gate threshold voltage(VSGT) of the PMOS transistor 112, which is typically in the range2V-4V. Additionally, in this scenario, VSD is essentially zero becausethe PMOS transistor 112 is in a conducting state. The voltage level atthe source node of the PMOS transistor 112 is nearly entirely droppedacross the diode stack 114 as the sum of the voltage drop across thediode stack 114 and the source/drain nodes of the PMOS transistor 112 isabout equal to the voltage at the source node of the PMOS transistor 112(e.g., VSD=VS−Vdiode˜45.6V−45.6V˜0V). In this manner, the PMOStransistor 112 is a low impedance or highly conducting component whenthe drive voltage level exceeds the knee voltage of the diode stack 114.

In contrast, the PMOS transistor 112 transitions to a high impedance orlow conductivity component when the difference between the drive voltagelevel of the current regulator 110 and VDC1 is reduced to less than thegate threshold voltage (VGST) of the PMOS transistor 112. Morespecifically, when the difference between the drive voltage level of thecurrent regulator 110 and the gate voltage (VSG) is below the gatethreshold voltage and decreasing, the PMOS transistor 112 changes fromthe low impedance “on” state to a closed-loop loop regulated state. Thisis the case when the current regulator 110 is adjusting to a low currentsetting and performs or implements the adjustment by reducing its outputvoltage.

The current regulator 110, the PMOS transistor 112 and the diode stack114 are in series electrically and conduct the same amount of current. A“low” current sourced by the current regulator 110 reduces the voltageover the diode stack 114, even below the knee voltage of the diode stack114. Below the gate threshold voltage, the impedance of the PMOStransistor 112 increases and therefore VSD increases and is additive tothe voltage dropped over the diode stack 114. This acts as closed-loopfeedback. To reduce the current, the current regulator 110 drops itsoutput voltage; this reduces VSG which results in an increase in VSD.This again reduces the voltage drop over the diode stack 114. An endcase would be a 0 A request, which reduces the output voltage of thecurrent regulator 110 to VDC1 and VSG to 0 V. In this case, theimpedance between the source and drain will be high and VSD voltage willbe approximately VDC1. At this point there will be no or near zerocurrent through the diode stack 114, and little to no voltage drop willoccur over the diode stack 114. In this manner, the PMOS transistor 112transitions to a high impedance component when the difference betweenthe drive voltage level of the current regulator 110 and VDC1 is reducedto less than the gate threshold voltage of the PMOS transistor 112. Theself-regulating nature of the PMOS transistor 112 as connected as shownin FIG. 2 is further demonstrated in FIG. 5.

FIG. 5 shows a chart diagram 500 of certain node voltages of the diodelaser system 100 of FIG. 2. In particular, a first characteristic 502 asshown in FIG. 5 describes the voltage characteristic of the diode stack114 as a function of drive current provided by the current regulator110. This characteristic is similar to the diode characteristic 402discussed above in connection with FIG. 4. A second characteristic 504as shown in FIG. 5 describes the voltage characteristic of VSG of thePMOS transistor 112 as a function of drive current provided by thecurrent regulator 110. VSG is initially zero and then increases to VSGTas the drive voltage level of the current regulator 110 approachesVDC1+VGST. A third characteristic 506 as shown in FIG. 5 describes thevoltage characteristic of VSD of the PMOS transistor 112 as a functionof drive current provided by the current regulator 110. VSD is initiallyat VDC1 and then is essentially zero when the drive voltage level of thecurrent regulator 110 exceeds VDC1+VGST. The slope in VSD at and beyondthis threshold is essentially zero due to the high gain of the PMOStransistor 112.

To a good approximation, the PMOS transistor 112 is a perfect or idealMOSFET, meaning there is no current flow between either one of thegate/source nodes and the gate/drain nodes of the PMOS transistor 112under normal operating conditions. Therefore, drive current output bythe current regulator 110 is equal to the current that flows through thePMOS transistor 112 and the diode stack 114.

When VSG>VSGT, the PMOS transistor 112 exhibits a low impedance and ishighly conductive. In this scenario, the PMOS transistor 112 does notcontribute to the electrical load. The electrical load is essentiallythe diode stack 114, and the current regulator 110 regulates the drivecurrent as intended.

According to the principles of the present disclosure, VK>(VDC1+VSGT).In this condition, when either increasing or decreasing drive current,the current regulator 110 regulates the current through the diode stack114, as intended. Also, when the drive voltage level is below VDC1+VSGT,there is impedance between the source/drain nodes of the PMOS transistor112 and thus any current would heat the PMOS transistor 112. Therefore,to ensure high-current operation occurs when the PMOS transistor 112exhibits a low impedance, VDC1 is selected to be less than (VK−VSGT).

Consider the case when the diode laser system 100 transitions from highoptical output to no or zero optical output. First, the drive currentsetpoint 124 is reduced. The current regulator 110 responds by loweringthe drive voltage and thereby the drive current. The drive voltage levelwill transition through VK and continue decreasing. When the drivevoltage level transitions below VDC1+VSGT, the impedance between thesource/drain nodes of the PMOS transistor 112 starts to increase (seeFIG. 5) and the PMOS transistor 112 starts to contribute to theelectrical load. As the current regulator 110 continues to reduce thedrive voltage, the voltage between the source/gate nodes of the PMOStransistor 112 tends towards zero and the impedance between thesource/drain nodes of the PMOS transistor 112 increases. Further, thevoltage between the source/drain nodes of the PMOS transistor 112 tendstowards VDC1 and therefore the voltage drop across the diode stack 114tends towards zero. This is illustrated in FIG. 5. Throughout, therelationship (VDC1+VSG)=(VDiode+VSD) holds true.

The arrangement shown in FIG. 2 (and FIG. 3) enables the currentregulator 110 to regulate current over the entire range between zero andthe rated current maximum for the diode stack 114. In the low currentcondition, diode load changes dramatically with changing current, but iscompensated by the changing load of the PMOS transistor 112. The currentregulator 110 adjusts the relatively small voltage across VSG and thePMOS transistor 112 matches this with a larger voltage adjustment acrossVSD. In other words, the current regulator 110 regulates drive currentby adjusting its output voltage along the VSG curve of FIG. 5. Inadjusting the output voltage between VDC1 and VDC1+VDC2, the voltagedrop across the diode stack 114 is changed between 0V and VDC1+VK. Theoutput voltage of the current regulator 110 tracks along the VSG curve,adjusting voltage to achieve and maintain the set drive current. Inadjusting the output voltage of the current regulator 110 aboveVDC1+VSGT, the voltage drop across the diode stack 114 changesidentically, volt-for-volt.

FIG. 6 is a circuit diagram of an example embodiment of the currentregulator 110 of the diode laser system 100 of FIG. 2. In general, thecurrent regulator 110 is a synchronous switching converter thatcomprises a controller 602, a high-side switch 604, a low-side switch606, a current feedback component 608, a diode 610, an output inductor612, a shunt resistor 614, and an output capacitor 616. Since thecurrent regulator 110 is incorporated into a high-power system orapplication (i.e., diode laser system 100), as opposed to a low-powersystem or application in which the synchronous converter topology istypically leveraged, a fast transient current may occur when the drivecurrent setpoint 124 is changed or when the diode laser system 100 isenabled or turned-on. That transient current may lead to significantcurrent overshoot, a transient oscillation, or an instable outputcurrent, all of which may be due to parasitic output wire inductance andother parasitic effects that together with the inductance andcapacitance presented by the output inductor 612 and output capacitor616, respectively, create a parasitic oscillator circuit.

To compensate, a damping resistor 618 is connected in series with theoutput capacitor 616, and one of skill will appreciate that the dampingresistor 618 may be utilized in a similar manner with respect to thecurrent regulator 110 of the diode laser system 100 of FIG. 3. Ingeneral, such a modification is contrary to conventional wisdom sincethe introduction of an additional resistance as contemplated may lead toamong other things additional power dissipation and heat generation dueto the high-power nature of the diode laser system 100. However, thediode laser system 100 is powered via mains supply (see FIG. 1), andfurther the electronics of the diode laser system 100 is in generalwater-cooled. In this manner, the cost of the trade-off betweenperformance and additional power dissipation and heat generation isminimized. FIG. 7 is a chart diagram of switching responses of thecurrent regulator 110 as a function of damping resistance.

In particular, a first characteristic 702 provided in FIG. 7 shows apronounced ringing in the switching response of the current regulator110. The value of the damping resistor 618 used in the bench test togenerate the first characteristic 702 is 0 (zero) ohms. Thus, the firstcharacteristic 702 captures the response of the current regulator 110with the damping resistor 618 omitted from the circuit. In contrast, asecond characteristic 704 provided in FIG. 7 shows a less-pronouncedringing in the switching response of the current regulator 110. Thevalue of the damping resistor 618 used in the bench test to generate thesecond characteristic 704 is 15 milliohms, all other variables beingheld equal. Additionally, a third characteristic 706 provided in FIG. 7shows an even less-pronounced ringing in the switching response of thecurrent regulator 110. The value of the damping resistor 618 used in thebench test to generate the third characteristic 706 is 100 milliohms,all other variables being held equal. Thus, both the secondcharacteristic 704 and the third characteristic 706 capture the responseof the current regulator 110 with the damping resistor 618 included inthe circuit, whereby the magnitude of ringing in the switching responseis inversely proportional to the value of the damping resistor 618.

FIG. 8 is a block diagram of a power scaling aspect according to thedisclosure. More specifically, an unspecified integer number of modulesof the current driver 104 a-n of the present disclosure are shownconnected together in parallel to drive a diode load 802 that comprisesa diode stack 804. The diode stack 114 comprises an unspecified integernumber of series-connected laser diodes. In this manner, multipleinstances of the current driver 104 may be coupled together in order todrive any particular diode load. Additionally, or alternatively, one orboth of VDC1 and VDC2 may be tuned independently of each other in orderto shift the operating point of a single instance of the current driver104 on an application- or implementation-specific basis. In someinstances, a non-complex clamp circuit may be built around the PMOStransistor 112 to ensure that VSG does not exceed an allowable orpermissible value that is a function of the type and rating of the PMOStransistor 112.

As shown in the figures and described above, the architecture of thecurrent driver 104 as contemplated is both modular and minimally complexwith respect to the number of components and connections. The currentdriver 104 is modular because the same may be used to drive manydifferent diode loads with as little as changing the magnitude ofvoltage bias to shift the operating point of the current driver 104. Thearchitecture of the current driver 104 does not need to change. Forexample, some or all components of the current driver 104 may exhibit acompliance voltage of 60V or less, which is advantageous becausehigh-voltage PCB board certification or verification is not required,and the current driver 104 may regulate at or more than 200 A through adiode load and be fast-modulating (e.g., 0-100 kHz, 0-100% duty cycle).A DC offset voltage may be leveraged to compensate for the turn-on orknee voltage of the diode load, and as such the current driver 104 maydrive additional diodes in series with the diode load at 100V, forexample, at the same current level even with a change in diode load andaccording to specification. This is despite the 100V voltage level beinggreater than the 60V compliance voltage of the current driver 104.Additionally, a power scaling aspect may be realized by connecting inparallel identical modules of the current driver 104. Further, thecurrent driver 104 may be minimally complex with respect to the numberof components and connections because a particular component of thecurrent driver 104, the switch element 112, may be realized with only asingle on-board (i.e., PCB) circuit component or element, namely a PMOSor NMOS transistor. The power supply that provides the DC offset voltagemay not be an on-board circuit component or element. The same is true ofthe diode load. It is contemplated that the current driver 104 may berealized in other ways, whereby other circuitry may be built into thecurrent driver 104. While additional development and component costswould be incurred and additional failure mechanisms may manifest, insome circumstances, such alternatives may be beneficial.

In light of such and other benefits and advantages, a current driver fora diode laser system is contemplated whereby the current driver includesor comprises a current regulator and a switch element. The currentregulator and the switch element may be configured and/or arrangedsimilar to the current regulator 110 and the switch element 112,respectively, as shown and described above in connection with at leastFIG. 2 and FIG. 3. In particular, the current regulator may beconfigured to receive from a power supply a bias voltage, where the biasvoltage shifts a compliance voltage range of the current regulator toinclude a knee voltage of a diode load, and where the bias voltage has avalue that corresponds to a lower bound of the shifted compliancevoltage range. An example of such a shifted compliance voltage range isillustrated and described above in connection with FIG. 4.

Additionally, the switch element may be coupled to the currentregulator, and may be configured to detect a voltage level that isoutput by the current regulator to supply drive current to the diodeload, where the switch element has (a) a high impedance when the voltagelevel that is output by the current regulator to supply drive current tothe diode load has a level that is less than the value of the biasvoltage and (b) a low impedance when the voltage that is output by thecurrent regulator to supply drive current to the diode load has a levelthat is greater than a sum of a turn-on threshold voltage of the switchelement and the value of the bias voltage. In some embodiments, theswitch element is in a transition state between the high impedance stateand the low impedance state when the voltage level that is output by thecurrent regulator to supply drive current to the diode load has a levelthat is greater than the value of the bias voltage but less than the sumof the turn-on threshold voltage of the switch element and the value ofthe bias voltage.

An example of such a variable-impedance switch element is a PMOStransistor, or an NMOS transistor, that when incorporated into a circuitarchitecture in a manner as shown and described above in connection withFIG. 2 and FIG. 3 exhibits I-V characteristics similar to that shown anddescribed above in connection with FIG. 4 and FIG. 5. Further, one ofskill will appreciate that in practice the level of particular voltagesas discussed throughout hold meaning when measured or defined withrespect to a particular circuit node, such as the gate node of thetransistor as shown and described above in connection with FIG. 2 andFIG. 3 for example. Other examples are possible.

In some embodiments, the compliance voltage range (and similarly theshifted compliance voltage range) is 60V or less in magnitude. Ingeneral, the compliance voltage range is defined by the compliancevoltage of the current regulator. In some embodiments, the currentregulator is powered by the difference between a supply voltage that isless than or equal to twice the bias voltage in magnitude and the biasvoltage (e.g., bias voltage=48V; supply voltage=96V or bias voltage=96V;supply voltage=144V). In some embodiments, a damping resistor is coupledin series with a capacitor that is coupled to an output node of thecurrent regulator, where the damping resistor has electrical resistanceselected from a range of ohmic values between 15 milliohms to 1000milliohms inclusive. An example of such a feature(s) is shown anddescribed above in connection with FIG. 6 and FIG. 7, where the currentregulator is a synchronous buck converter that exhibits a modulationfrequency of about 100 kHz or less, or equivalently can be switched onthe order of 10 μs. A particular ohmic value or electrical resistancevalue of the damping resistor may be selected on animplementation-specific basis, so as to tune signal response in the timedomain to minimize ringing, exhibit particular rise/fall times, andetc., as desired.

Although only certain exemplary embodiments have been described indetail above, those skilled in the art will readily appreciate that manymodifications are possible in the exemplary embodiments withoutmaterially departing from the novel teachings and advantages of thisdisclosure. Aspects of embodiments disclosed above can be combined inother combinations to form additional embodiments. All suchmodifications are intended to be included within the scope of thistechnology.

What is claimed is:
 1. A current driver for a diode system, comprising acurrent regulator that is configured to receive from a power supply abias voltage, wherein the bias voltage shifts a compliance voltage rangeof the current regulator to include a knee voltage of a diode load, andwherein the bias voltage has a value that corresponds to a lower boundof the shifted compliance voltage range; and a switch element that iscoupled to the current regulator and that is configured to detect avoltage level that is output by the current regulator to supply drivecurrent to the diode load, wherein the switch element has (a) a highimpedance when the voltage level that is output by the current regulatorto supply drive current to the diode load is less than the value of thebias voltage and (b) a low impedance when the voltage level that isoutput by the current regulator to supply drive current to the diodeload is greater than a sum of a threshold voltage value and the value ofthe bias voltage.
 2. The current driver of claim 1, wherein thecompliance voltage range is 60V or less in magnitude.
 3. The currentdriver of claim 1, wherein the switch element is a PMOS transistor, anda source node of the PMOS transistor is coupled to the currentregulator, a gate node of the PMOS transistor is biased by the biasvoltage, and a drain node of the PMOS transistor provides for drivecurrent a path to the diode load.
 4. The current driver of claim 1,wherein the switch element is an NMOS transistor, and a source node ofthe NMOS transistor is coupled to the current regulator, a gate node ofthe NMOS transistor is biased by the bias voltage, and a drain node ofthe NMOS transistor provides for drive current a path to the diode load.5. The current driver of claim 1, wherein the current regulator ispowered by a difference between a supply voltage that is twice the biasvoltage in magnitude and the bias voltage.
 6. The current driver ofclaim 1, wherein the current regulator is powered by a differencebetween a supply voltage that is less than twice the bias voltage inmagnitude and the bias voltage.
 7. The current driver of claim 1,further comprising a damping resistor that is coupled in series with acapacitor that is coupled to an output node of the current regulator,wherein the damping resistor has an electrical resistance selected frombetween 15 milliohms to 1000 milliohms inclusive.
 8. The current driverof claim 1, wherein the current regulator is a synchronous buckconverter.
 9. The current driver of claim 1, wherein a modulationfrequency of the current regulator is about 100 kHz or less.
 10. A diodesystem, comprising: a diode load; a power supply that is configured tooutput a bias voltage; a current regulator that is coupled to the powersupply and that is configured to receive from the power supply the biasvoltage, wherein the bias voltage shifts a compliance voltage range ofthe current regulator to include a knee voltage of the diode load, andwherein the bias voltage has a value that corresponds to a lower boundof the shifted compliance voltage range; and a switch element that iscoupled to the current regulator and that is configured to detect avoltage level that is output by the current regulator to supply drivecurrent to the diode load, wherein the switch element has (a) a highimpedance when the voltage level that is output by the current regulatorto supply drive current to the diode load is less than the value of thebias voltage and (b) a low impedance when the voltage level that isoutput by the current regulator to supply drive current to the diodeload is greater than a sum of a threshold voltage value and the value ofthe bias voltage.
 11. The diode system of claim 10, wherein thecompliance voltage range is 60V or less in magnitude.
 12. The diodesystem of claim 10, wherein the switch element is a PMOS transistor, anda source node of the PMOS transistor is coupled to the currentregulator, a gate node of the PMOS transistor is biased by the biasvoltage, and a drain node of the PMOS transistor provides for drivecurrent a path to the diode load.
 13. The diode system of claim 10,wherein the switch element is an NMOS transistor, and a source node ofthe NMOS transistor is coupled to the current regulator, a gate node ofthe NMOS transistor is biased by the bias voltage, and a drain node ofthe NMOS transistor provides for drive current a path to the diode load.14. The diode system of claim 10, wherein the current regulator ispowered by a difference between a supply voltage that is less than orequal to twice the bias voltage in magnitude and the bias voltage. 15.The diode system of claim 10, wherein the diode load includes aplurality of series-connected diodes.
 16. The diode system of claim 10,further comprising a damping resistor that is coupled in series with acapacitor that is coupled to an output node of the current regulator,wherein the damping resistor has an electrical resistance selected frombetween 15 milliohms to 1000 milliohms inclusive.
 17. The diode systemof claim 10, wherein the current regulator is a synchronous buckconverter.
 18. The diode system of claim 10, wherein a modulationfrequency of the current regulator is about 100 kHz or less.
 19. Thediode system of claim 10, wherein the power supply includes a pair ofseries-connected voltage sources, and wherein a first voltage source ofthe pair is configured to output the bias voltage and a second voltagesource of the pair is configured to output a supply voltage that has amagnitude equal to or different than a magnitude of the bias voltage.20. A current driver for a diode laser system, comprising a currentregulator that is configured to receive from a power supply a biasvoltage, wherein the bias voltage shifts a compliance voltage range ofthe current regulator to include a knee voltage of a laser diode load,and wherein the bias voltage has a value that corresponds to a lowerbound of the shifted compliance voltage range; and a transistor that iscoupled to the current regulator and that is configured to detect avoltage level that is output by the current regulator to supply drivecurrent to the laser diode load, wherein the transistor has (a) a highimpedance when the voltage level that is output by the current regulatorto supply drive current to the laser diode load is less than the valueof the bias voltage and (b) a low impedance when the voltage level thatis output by the current regulator to supply drive current to the laserdiode load is greater than a sum of a threshold voltage value and thevalue of the bias voltage; wherein a source node of the transistor iscoupled to the current regulator, a gate node of the transistor isbiased by the bias voltage, and a drain node of the transistor providesfor drive current a path to the laser diode load.
 21. A method fordriving a diode load, comprising: by a diode system: outputting, by acurrent regulator of the diode system, a voltage level to supply drivecurrent to a diode load of the diode system, wherein a bias voltagesupplied to the current regulator by a power supply corresponds to alower bound of a compliance voltage range of the current regulator, andwherein the compliance voltage range includes a knee voltage of thediode load; and adjusting, by a switch element of the diode system, animpedance of the switch element to (a) a higher impedance when thevoltage level that is output by the current regulator to supply drivecurrent to the diode load transitions from a sum of a threshold voltagevalue associated with the switch element and the value of the biasvoltage towards the value of the bias voltage and (b) a lower impedancewhen the voltage level that is output by the current regulator to supplydrive current to the diode load transitions from the value of the biasvoltage towards the sum of a threshold voltage value associated with theswitch element and the value of the bias voltage.